1. Field of the Invention
The present invention pertains generally to photovoltaic system charge controllers and, more particularly, to modulation controls for power converters in photovoltaic system charge controllers, particularly high voltage maximum power point tracking photovoltaic system charge controllers.
2. Brief Discussion of the Related Art
Photovoltaic (PV) systems that produce electricity from solar energy have established themselves as a successful and reliable option for electrical power generation. Photovoltaic systems have continually been gaining in popularity as the cost of such systems has been reduced, as the cost of utility-supplied power has escalated and as greater attention has been paid to the need for safe, renewable, alternative energy sources. Basically, a photovoltaic system includes a photovoltaic (PV) array made up of one or more PV panels or modules composed of photovoltaic cells capable of converting solar energy into direct current (DC) electrical energy, a battery bank made up of one or more batteries for storing the electrical energy produced by the photovoltaic array, and a charge controller for controlling the charging of the one or more batteries with the electrical energy produced by the photovoltaic array. The direct current (DC) electrical energy produced by the photovoltaic array and/or stored in the battery bank is available to power a DC load. In some systems, the DC load may include an inverter used to convert the direct current (DC) electrical energy into alternating current (AC) electrical energy suitable to power AC loads. Photovoltaic systems are sometimes employed to power loads independently of utility power, such as where electrical power from the public utility grid is unavailable or not feasible, and these photovoltaic systems are commonly referred to as “off-grid” and “stand-alone” photovoltaic systems. In other instances, photovoltaic systems known as “on-grid” and “grid-connected” photovoltaic systems are employed to supply electrical power to the public utility grid as explained further below.
In accordance with programs commonly referred to as “net metering”, many public utilities provide compensation for the net electrical power that is fed into the utility grid from grid-connected photovoltaic systems. The electrical power produced by grid-connected photovoltaic systems typically is used first to operate any connected end load, such as various conventional electrical appliances and devices, and the excess electrical power not consumed by the connected end load is then supplied to the utility grid. If the photovoltaic system fails to produce enough electrical power to operate the connected end load, electricity is drawn from the utility grid to power the load. Through net metering programs, the owner of the grid-connected photovoltaic system is compensated for the net outflow of electrical power from the photovoltaic system into the utility grid.
Grid-connected photovoltaic systems utilize inverters, conventionally referred to as “on-grid” or “grid-connected” inverters, that transform the direct current (DC) electrical power produced by the photovoltaic system into alternating current (AC) electrical power suitable for being supplied to the utility grid and for powering any other connected AC end load. Grid-connected inverters normally function to ensure that the AC electrical power supplied to the utility grid is in sinusoidal form, synchronized to the frequency of the grid, and limited to a feed voltage, i.e. the output voltage of the inverter, that is no higher than the grid voltage. One way in which the AC electrical power output from an on-grid inverter can be supplied to the utility grid and/or another connected AC end load involves connecting the inverter output to an electrical distribution panel as typically found in residential, commercial, business and/or other types of buildings or structures. The source of DC electrical input to the on-grid inverter may come from various sources including electrical energy stored in the battery bank of the photovoltaic system, flywheels and/or fuel cells, for example.
Photovoltaic systems have been designed with traditional charge controllers that do not employ maximum power point tracking (MPPT), and such charge controllers may be referred to as non-MPPT charge controllers. Non-MPPT charge controllers connect the PV array directly to the battery bank for charging. Usually there is a mismatch between the output voltage of the PV array and the voltage required to charge the battery bank that results in under-utilization of the maximum power output from the PV array. The reason for the mismatch is that most PV modules are rated to produce a nominal 12V under standard test conditions but, because they are designed for worse than standard test conditions, in actual fact they produce significantly more voltage. On the other hand, a nominal 12V battery requires close to an actual 12V (14V typically) depending on battery state of charge. When a non-MPPT charge controller is charging the battery, the PV module is frequently forced to operate at a battery voltage that is less than the optimal operating voltage at which the PV module is capable of producing its maximum power. Hence, non-MPPT charge controllers artificially limit power production to a sub-optimal level by constraining the PV array from operating at maximum output power.
Maximum power point tracking (MPPT) charge controllers address the aforesaid disadvantage of non-MPPT charge controllers by managing the voltage mismatch between the PV array and the battery bank through the use of power electronics. The primary functions performed by MPPT charge controllers involve measuring the PV module output to find the maximum power voltage (Vmp), i.e. the voltage at which the PV module is able to produce maximum power, operating the PV module at the maximum power voltage to extract or harvest full power (watts) from the PV array, regardless of the present battery voltage (VB), and protecting the battery from overcharge.
Photovoltaic modules are made up of photovoltaic (PV) cells that have a single operating point where the values of the current (I) and voltage (V) of the cell result in a maximum power output. The maximum power voltage Vmp varies with operating conditions including weather, sunlight intensity, shading, and PV cell temperature. As the maximum power voltage Vmp of the PV module varies, MPPT charge controllers “track” the Vmp and adjust the ratio between the maximum power voltage and the current delivered to the battery in order to match what the battery requires. MPPT charge controllers utilize a control circuit or logic to search for the maximum power output operating point and employ power electronics to extract the maximum power available from a PV module.
MPPT charge controllers generally employ power converters designed for a higher input voltage than output voltage, hence Vmp>VB. The power converters are conventionally designed to include a DC to DC converter that receives the maximum power voltage Vmp from the PV array as converter input and converts the maximum power voltage to battery voltage VB as converter output. An increase in battery charge current is realized by harvesting PV module power that would be left unharvested using a non-MPPT charge controller. As the maximum power voltage varies, the actual charge current increase that is realized will likewise vary. Generally speaking, the greater the mismatch or disparity between the PV array maximum power voltage Vmp and the battery voltage VB, the greater the charge current increase will be. The charge current increase will ordinarily be greater in cooler temperatures because the available power output and the maximum power voltage of the PV module increase as the photovoltaic cell temperature decreases. In addition, lower battery voltage, as in the case of a highly discharged battery, will result in a greater charge current increase.
Most MPPT charge controllers utilize power electronics designed to include a “buck” converter having topology to “buck” or “step-down” a higher input voltage to a lower output voltage. Buck converters, also known as “step-down” converters, are familiar in the field of power electronics and essentially include an inductor and two complementary switches to achieve unidirectional power flow from input to output. A first of the switches is ordinarily a controlled switch such as a MOSFET (metal oxide semiconductor field effect transistor) or other transistor, and the second of the switches is ordinarily an uncontrolled switch such as a discrete power diode. The buck converter alternates between connecting the inductor to the input voltage (VA) from the PV array to store energy in the inductor and discharging the inductor into the battery bank. When the first switch is turned “on” for a time duration, the second switch becomes reverse biased and the inductor is connected to the input voltage VA. There is a positive voltage (VL) across the inductor equal to the input voltage VA minus the output voltage VB, hence VL=VA−VB, and there is an increase in the inductor current (IL). In this “on” state, energy is stored in the inductor. When the first switch is turned “off”, inductor current IL continues to flow due to the inductor energy storage, resulting in a negative voltage across the inductor (VL=−VB). The inductor current now flows through the second switch, which is forward biased, and current IL through the inductor decreases. In this “off” state, energy continues to be delivered to the output until the first switch is again turned “on” to begin another on-off cycle. The buck converter is operated in continuous conduction mode (CCM) when the current through the inductor never goes to zero during the commutation cycle. The buck converter is operated in discontinuous conduction mode (DCM) when the current through the inductor goes to zero every commutation cycle.
In addition to voltage stepping-down applications, DC to DC converters have been used in the past to “boost” or “step-up” a lower input voltage to a higher output voltage. These types of DC to DC converters are commonly referred to as “boost” or “step-up” converters.
Some of the limitations to using a buck converter in MPPT charge controllers for photovoltaic systems include high peak currents and voltages with attendant high power losses, and increasing control problems as the input voltage increases. The efficiency of buck converters can be improved to some extent using a technique known as “synchronous rectification”. In synchronous rectification, the discrete power diode that serves as the second switch in the buck converter can be replaced with a MOSFET which, like all power MOSFETs, has an intrinsic or inherent anti-parallel parasitic body diode between the source and the drain of the MOSFET's transistor. When the body diode of the MOSFET of the second switch is forward biased and conducting current, the transistor of the MOSFET of the second switch is turned “on” a short time after its body diode has started to conduct. The transistor of the MOSFET of the second switch is turned “off” a short time before the MOSFET of the first switch in the buck converter is going to turn back “on”. The MOSFET of the second switch in the “on” state behaves as a low value resistance, reducing the forward voltage and yielding lower losses. While this MOSFET is “on”, the forward voltage drop of the body diode is limited to the “on” resistance of its transistor. This forward voltage drop can be significantly lower than the voltage drop in the discrete power diode referred to above as the second switch in the buck converter, thereby lowering conduction losses.
Sophisticated MPPT photovoltaic system charge controllers send out a series of short charging pulses to the battery bank. The controllers monitor the state of the battery bank and adjust the pulses as needed to regulate the amount of charge sent to the battery bank. This technique is commonly referred to as “pulse width modulation,” i.e. PWM. Based on the monitored system parameters, the controllers generate commands representing required duty cycles for the power converters of the controllers, which result in the appropriate switching signals being applied to the switches of the power converters. Accordingly, power flow through the converters and electrical output from the converters are controlled in accordance with a modulation control scheme executed by the controllers.
Most conventionally available photovoltaic system charge controllers that utilize a buck converter to implement maximum power point tracking (MPPT) are limited to an input of 150V, one exception being the MPPT charge controller developed by Australian Energy Research Laboratory (AERL) which is capable of handling an input of 250V. Conventional on-grid inverters, however, operate with high voltage PV arrays up to 600V, such that presently available MPPT charge controllers for photovoltaic systems are generally unsuitable for use in grid-connected photovoltaic systems due to their inability to handle the high voltage.
A high voltage (HV) bidirectional maximum power point tracking (MPPT) charge controller that can be used in photovoltaic systems having a high voltage photovoltaic array of up to 600V is the subject of U.S. patent application Ser. No. 12/896,427 filed Oct. 1, 2010, which is commonly owned by the Assignee of the subject patent application and the entire disclosure of which is incorporated herein by reference. A high voltage bidirectional maximum power point tracking charge controller described in the aforementioned prior application incorporates a series-connected dual active bridge (DAB) bidirectional DC to DC converter that utilizes MOSFETs as the switches in each bridge. The bidirectional DC to DC converter receives DC input from the photovoltaic array and operates in a first direction of power flow to step-down the voltage of the DC input received from the photovoltaic array to obtain a stepped-down DC output of appropriate voltage to optimally charge the battery bank. The bidirectional DC to DC converter also receives DC input from the battery bank and operates in a second or reverse direction of power flow to step-up the voltage of the DC input received from the battery bank to obtain a stepped-up DC output of appropriate voltage for a high voltage DC load, which can be an inverter for transforming DC electricity received from the charge controller into AC electricity appropriate for being supplied to a public utility grid and/or to another connected AC end load.
The aforesaid dual active bridge (DAB) bidirectional DC to DC converter employs a primary bridge having four MOSFETs as switches, a secondary bridge having four MOSFETs as switches, and a transformer electrically connecting the primary and secondary bridges. When the converter is operated in the first direction of power flow, the primary bridge receives DC input from the PV array and the secondary bridge supplies the stepped-down DC output to the battery bank in the manner of a buck converter. Conversely, when the converter is operated in the second direction of power flow, the secondary bridge, which now functions as the primary bridge, receives DC input from the battery bank, and the primary bridge, which now functions as the secondary bridge, supplies the stepped-up DC output to the high voltage DC load in the manner of a boost converter. Operation of the converter involves turning the MOSFETs on and off by controlling the electrical switching signals applied to the gates of the MOSFETs in accordance with a modulation control scheme to control the power flow through the converter and the electrical output from the converter.
Dual active bridge (DAB) bidirectional DC to DC converters that utilize the transformer's leakage inductance Llk, or the leakage inductance Llk and another inductance connected in series with any of the transformer's terminals, to control the converter's bidirectional power flow while allowing each switch in the bridges to be implemented as a MOSFET have previously been considered by DeDoncker et al in U.S. Pat. No. 5,027,264; by DeDoncker et al in “A Three-phase Soft-Switched High Power Density DC/DC Converter For High Power Applications” (1988 IEEE); by DeDoncker et al in “A Three-Phase Soft-Switched High-Power-Density dc/dc Converter for High-Power Applications” (1991 IEEE); by Kheraluwala et al in “Performance Characterization of a High-Power Dual Active Bridge dc-to-dc Converter” (1992 IEEE); by Vangen et al in “Dual Active Bridge Converter With Large Soft-Switching Range” (1993 The European Power Electronic Association); by Vangen et al in “Soft-Switched High-Frequency, High-Power DC/AC Converter With IGBT” (1992 IEEE); by Vangen et al in “Efficient High-Frequency Soft-Switched Power Converter With Signal Processor Control” (1991 IEEE); by Schibli in “Symmetrical Multilevel Converters With Two Quadrant DC-DC Feeding” (2000 Ecole Polytechnique Federale de Lausanne); by Song et al in “A New Soft Switching Technique for Bi-directional Power Flow, Full-Bridge DC-DC Converter” (2002 IEEE); by Chan et al in “A Phase-Shift Controlled Bi-directional DC-DC Converter” (1999 IEEE); and by Chan et al in “ZCS-ZVS bi-directional phase-shifted DC-DC converter with extended load range” (2003 IEEE). These converters avoid the drawback of having to fight the leakage inductance as in the buck-derived isolated bridge converters considered by Sabaté et al in “Design Considerations For High-Voltage High-Power Full-Bridge Zero-Voltage-Switched PWM Converter” (1990 IEEE); by Cho et al in “Novel Full Bridge Zero-Voltage-Transition PWM DC/DC Converter for High Power Applications” (1994 IEEE); by Cho et al in “Zero-Voltage and Zero-Current-Switching Full Bridge PWM Converter for High Power Applications” (1994 IEEE); and by Cuadros et al in “Design Procedure and Modeling of High Power, High Performance, Zero-Voltage Zero-Current Switched, Full-Bridge PWM Converter” (1997 IEEE). The switches in the primary and secondary bridges of the dual active bridge (DAB) bidirectional DC to DC converters are controlled to produce phase-shifted square waves across the transformer (or across the series connection of a transformer and an inductance). This type of modulation control may be referred to as “phase-shift control.” U.S. Pat. No. 5,027,264 demonstrates that, if MOSFETs are used as the switches in each bridge, the dual active bridge converter can operate both with zero-voltage switching (ZVS) and synchronous rectification within a constrained range, i.e. when the voltage conversion ratio is close to the transformer's turn ratio. Within this constrained range, the dual active bridge converter can exhibit high efficiency at high switching frequencies by avoiding power losses and electromagnetic noise associated with diode reverse recovery characteristics under hard switching. Hard switching forces the diode to turn off by turning on the opposite switch in the same leg of the bridge, which produces a high negative peak current in the diode as well as discharging the node capacitance through the switch itself. Consequently, switch power losses are significantly increased. The MOSFET's intrinsic body diode exhibits poorer reverse recovery characteristics than a discrete power diode and, as a result, induces higher power losses than a discrete power diode when hard-switched off. Outside the constrained range, however, zero-voltage switching is lost. In particular, when the voltage conversion ratio is much lower or much higher than the transformer's turn ratio, some of the diodes get hard switched off. When the voltage conversion ratio is much lower than the transformer's turn ratio, zero-voltage switching is lost on the secondary bridge. When the voltage conversion ratio is much higher than the transformer's turn ratio, zero-voltage switching is lost on the primary bridge.
A modulation control for a dual active bridge DC to DC converter that extends its zero-voltage switching (ZVS) operating region and reduces its root mean square (RMS) current value in some regions of the converter's operating range when compared to the aforementioned standard phase-shift control is presented by Kheraluwala et al in “Performance Characterization of a High-Power Dual Active Bridge dc-to-dc Converter”, by Vangen et al in “Dual Active Bridge Converter with Large Soft-Switching Ranges”, and by Vangen et al in “Soft-Switched High-Frequency, High Power DC/AC converter with IGBT.” This modulation control approach employs two angles as the command variables, i.e. angle α that sets the phase-shift between the input-side bridge legs and angle φ that sets the phase-shift between the primary and secondary bridges. This type of modulation control, which may be referred to as “two-angle control”, generates a three-level voltage waveform between the nodes on the primary bridge but a two-level voltage waveform between the nodes on the secondary bridge. Adding levels to the bridge voltages in this manner expands the conditions under which zero-voltage switching can be attained. However, restrictions on the values for the normalized voltage conversion ratio preclude bidirectional operation under certain conditions.
In order to alleviate the power losses incurred in the dual active bridge DC to DC converter by operation under hard switching conditions, three-angle control sequences that can generate three-level voltage waveforms in both bridges, i.e. between the nodes in the primary bridge and also between the nodes in the secondary bridge, are proposed by Vangen et al in “Efficient High-Frequency Soft-Switched Power Converter With Signal Processor Control.” This approach to control modulation, which may be referred to as “three-angle control,” provides more possibilities for soft switching conditions. Switch commutation is performed with zero-voltage switching (ZVS) and zero-current switching (ZCS) for certain intervals of each switching semi-cycle. The three-angle control approach generates a rectangular mode, a triangular mode and a trapezoidal mode of operation. The trapezoidal mode is recommended by Schibli in “Symmetrical Multilevel Converters With Two-Quadrant DC-DC Feeding.” Schibli recommends this mode of operation for its specific case of high voltage and IGBT switch implementation. However, this type of trapezoidal mode with zero-current during a portion of the switching semi-cycle is disadvantageous because it unnecessarily increases RMS current value and prevents zero-voltage switching operation in a certain range.
In “A New Soft Switching Technique for Bi-directional Power Flow, Full-Bridge DC-DC Converter,” Song et al proposed a modulation control scheme for dual active bridge converter operation similar to that discussed by Schibli. Song et al only considers voltage conversion rate values of strictly less than 1, and bidirectional operation for the same circuit design with this control scheme is not possible. The control scheme proposed by Song et al can only assist the dual active bridge converter's efficiency in a very narrow range and for unidirectional operation.
A dual active bridge DC to DC converter formed with a half-bridge and a full-bridge, together with a switch control scheme similar to that proposed by Song et al, is discussed by Zhang et al in “An Improved Dual Active Bridge DC/DC Converter” (2001 IEEE). The analysis and experimental comparisons are demonstrated by Zhang et al for a voltage conversion ratio equal to 1, under which conditions a dual active bridge design having the phase-shift control proposed by DeDoncker et al always operates with zero-voltage switching and would exhibit superior performance compared to Zhang et al's converter.